Current-mode control for radio-frequency power amplifiers

ABSTRACT

Current-mode control for radio-frequency (RF) power amplifiers. In some embodiments, an RF power amplifier control circuit can include a sensor configured to measure a base current of a power amplifier and generate a sensed current. The control circuit can further include a sensing node configured to receive a reference current and perform a current-mode operation with the sensed current to yield an error current. The control circuit can further include a control loop configured to generate a control signal based on the error current to adjust an operating parameter of the power amplifier.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a divisional of U.S. application Ser. No. 13/967,329filed Aug. 14, 2013, entitled SYSTEMS, CIRCUITS AND METHODS RELATED TOCONTROLLERS FOR RADIO-FREQUENCY POWER AMPLIFIERS, which claims priorityto and the benefit of the filing date of U.S. Provisional ApplicationNo. 61/683,674 filed Aug. 15, 2012 and entitled CLAMPING CIRCUITS ANDMETHODS BASED ON BETA-TRACKING TEMPERATURE-COMPENSATED BASE CURRENT, thebenefits of the filing dates of which are hereby claimed and thedisclosures of which are hereby expressly incorporated by reference intheir entirety.

BACKGROUND

1. Field

The present disclosure generally relates to circuits and methods forcontrolling radio-frequency (RF) amplifiers, and more particularly, toclamping circuits and methods for such RF amplifiers.

2. Description of the Related Art

In wireless devices such as cellular phones, different components suchas a power amplifier (PA) draw current from a battery which typicallyhas a limited capacity. Such a PA can also generate heat if notcontrolled properly. Thus, it is desirable to have the PA provided witha limited current, so that the current drawn from the battery does notexceed some specified current limit, while maintaining the performanceof the PA.

SUMMARY

In accordance with a number of implementations, the present disclosurerelates to a radio-frequency power amplifier control circuit. Thecontrol circuit includes a first circuit configured to generate areplica base current from a base current provided to a power amplifier(PA), with the replica base current being representative of a collectorcurrent of the PA scaled by a beta parameter. The control circuitfurther includes a second circuit configured to generate a beta-trackingreference current from a temperature-compensated voltage and a baseresistance associated with the PA. The control circuit further includesa current steering circuit configured to receive the replica basecurrent and the beta-tracking reference current and generate aproportional current.

In some embodiments, the temperature-compensated voltage can include atemperature-compensated bandgap voltage.

In some embodiments, the current steering circuit can be configured togenerate the proportional current under a selected condition based onthe replica base current and the beta-tracking reference current. Thecontrol circuit can further include a PA base driver in communicationwith the current steering circuit to receive the proportional current.The current steering circuit can be configured to generate theproportional current by comparing a replica base voltage correspondingto the replica base current and a beta-tracking reference voltagecorresponding to the beta-tracking reference current. Each of thereplica base voltage and the beta-tracking reference voltage can begenerated by providing the respective current into a matched resistor.The selected condition can include the replica base voltage exceedingthe beta-tracking reference voltage.

In some embodiments, the base current can be measured by a fingersensor. The finger sensor can be part of a control loop that isconfigured to generate an error current based on comparison of the basecurrent and a ramp current. The comparison of the base current and theramp current can be performed in a current-mode. The error current canbe obtained by subtracting the base current from the ramp current. Thecontrol loop can further include a trans-impedance amplifier configuredto amplify the error current.

In some embodiments, the power amplifier can include a gallium arsenide(GaAs) heterojunction bipolar transistor (HBT) power amplifier.

In some embodiments, the control circuit can further include apre-charging system configured to pre-charge a selected node of thecontrol circuit. The pre-charging system can include a sensor circuitconfigured to generate a control signal based on comparison of a basevoltage of the PA and a reference voltage selected to be lower than athreshold voltage of the PA. The pre-charging system can further includean actuator circuit in communication with the sensor circuit, with theactuator circuit being configured to receive the control signal from thesensor circuit. The actuator circuit can be further configured to enableor disable, based on the control signal, a pre-charge current forpre-charging the selected node of the control circuit.

In some implementations, the present disclosure relates to a method forcontrolling a radio-frequency power amplifier. The method includesgenerating a replica base current from a base current provided to apower amplifier (PA), with the replica base current representative of acollector current of the PA scaled by a beta parameter. The methodfurther includes generating a beta-tracking reference current from atemperature-compensated voltage and a base resistance associated withthe PA. The method further includes generating a proportional currentbased on the replica base current and the beta-tracking referencecurrent.

In some embodiments, the method can further include providing theproportional current to a clamping node of a PA base driver under aselected condition.

According to a number of implementations, the present disclosure relatesto a radio-frequency (RF) module. The RF module includes a packagingsubstrate configured to receive a plurality of components. The RF modulefurther includes a power amplifier (PA) disposed over the packagingsubstrate. The RF module further includes a control circuit disposedover the packaging substrate and interconnected to the power amplifier.The control circuit includes a first circuit configured to generate areplica base current from a base current provided to the PA, with thereplica base current being representative of a collector current of thePA scaled by a beta parameter. The control circuit further includes asecond circuit configured to generate a beta-tracking reference currentfrom a temperature-compensated voltage and a base resistance associatedwith the PA. The control circuit further includes a current steeringcircuit configured to generate a proportional current based on thereplica base current and the beta-tracking reference current. The RFmodule further includes a plurality of connectors configured to provideelectrical connections between the power amplifier, the control circuit,and the packaging substrate.

In some embodiments, the power amplifier can be disposed on a first dieand the control circuit can be disposed on a second die, with each ofthe first die and the second die being mounted on the packagingsubstrate.

According to some teachings, the present disclosure relates to aradio-frequency (RF) device. The RF device includes a transceiverconfigured to generate an RF signal. The RF device further includes apower amplifier (PA) in communication with the transceiver, with the PAconfigured to amplify the RF signal. The RF device further includes acontrol circuit in communication with the PA. The control circuitincludes a first circuit configured to generate a replica base currentfrom a base current provided to the PA, with the replica base currentbeing representative of a collector current of the PA scaled by a betaparameter. The control circuit further includes a second circuitconfigured to generate a beta-tracking reference current from atemperature-compensated voltage and a base resistance associated withthe PA. The control circuit further includes a current steering circuitconfigured to generate a proportional current based on the replica basecurrent and the beta-tracking reference current. The RF device furtherincludes an antenna in communication with the PA, with the antenna beingconfigured to facilitate transmission of the amplified RF signal.

In some embodiments, the RF device can include a wireless device such asa cellular phone.

In a number of implementations, the present disclosure relates to aradio-frequency power amplifier control circuit. The control circuitincludes a sensor configured to measure a base current of a poweramplifier and generate a sensed current. The control circuit furtherincludes a sensing node configured to receive a reference current andperform a current-mode operation with the sensed current to yield anerror current. The control circuit further includes a control loopconfigured to generate a control signal based on the error current toadjust an operating parameter of the power amplifier.

In some embodiments, the power amplifier can include a gallium arsenide(GaAs) heterojunction bipolar transistor (HBT) power amplifier. The GaAsHBT power amplifier can be configured to operate in, for example,GSM/GPRS communication protocols.

In some embodiments, the sensor can include a finger sensor configuredto sense the base current of the power amplifier to yield the sensedcurrent. In some embodiments, the current-mode operation can includesubtracting the sensed current from the reference current to yield theerror current. In some embodiments, the power amplifier control circuitcan be substantially free of a sense resistor between the sensor and thesensing node. In some embodiments, the reference current can be obtainedfrom an external analog control voltage. In some embodiments, thesensing node can be configured to be regulated to substantially maintaina voltage that tracks a battery voltage. In some embodiments, thecontrol loop can include a trans-impedance amplifier configured toamplify the error current. In some embodiments, the control loop caninclude a closed control loop. In some embodiments, the power amplifiercontrol circuit can be substantially free of an external bypasscapacitor to thereby reduce cost and size associated with the poweramplifier circuit.

In some embodiments, the control circuit can further include apre-charging system configured to pre-charge a selected node of thecontrol circuit. The pre-charging system can include a sensor circuitconfigured to generate a control signal based on comparison of a basevoltage of the PA and a reference voltage selected to be lower than athreshold voltage of the PA. The pre-charging system can further includean actuator circuit in communication with the sensor circuit, with theactuator circuit being configured to receive the control signal from thesensor circuit. The actuator circuit can be further configured to enableor disable, based on the control signal, a pre-charge current forpre-charging the selected node of the control circuit.

According to some implementations, the present disclosure relates to amethod for controlling a radio-frequency power amplifier. The methodincludes measuring a base current of a power amplifier to generate asensed current. The method further includes performing a current-modeoperation between a reference current and the sensed current to yield anerror current. The method further includes generating a control signalbased on the error current. The method further includes adjusting anoperating parameter of the power amplifier based on the control signal.

In some implementations, the present disclosure relates to aradio-frequency (RF) module. The RF module includes a packagingsubstrate configured to receive a plurality of components. The RF modulefurther includes a power amplifier disposed over the packagingsubstrate. The RF module further includes a control circuit disposedover the packaging substrate and interconnected to the power amplifier.The control circuit includes a sensor configured to measure a basecurrent of the power amplifier and generate a sensed current. Thecontrol circuit further includes a sensing node configured to receive areference current and perform a current-mode operation with the sensedcurrent to yield an error current. The control circuit further includesa control loop configured to generate a control signal based on theerror current to adjust an operating parameter of the power amplifier.The RF module further includes a plurality of connectors configured toprovide electrical connections between the power amplifier, the controlcircuit, and the packaging substrate.

In some embodiments, the power amplifier can be disposed on a first dieand the control circuit can be disposed on a second die, with each ofthe first die and the second die being mounted on the packagingsubstrate.

In accordance with a number of implementations, the present disclosurerelates to a radio-frequency (RF) device. The RF device includes atransceiver configured to generate an RF signal. The RF device furtherincludes a power amplifier in communication with the transceiver, withthe power amplifier being configured to amplify the RF signal. The RFdevice further includes a control circuit in communication with thepower amplifier. The control circuit includes a sensor configured tomeasure a base current of the power amplifier and generate a sensedcurrent, a sensing node configured to receive a reference current andperform a current-mode operation with the sensed current to yield anerror current, and a control loop configured to generate a controlsignal based on the error current to adjust an operating parameter ofthe power amplifier. The RF device further includes an antenna incommunication with the power amplifier, with the antenna beingconfigured to facilitate transmission of the amplified RF signal.

In some embodiments, the RF device can include a wireless device. Such awireless device can be configured to operate as, for example, a GSM/GPRScommunication device.

In some implementations, the present disclosure relates to apre-charging system for a radio-frequency power amplifier controlcircuit. The pre-charging system includes a sensor circuit configured togenerate a control signal based on comparison of a base voltage of apower amplifier (PA) and a reference voltage selected to be lower than athreshold voltage of the PA. The pre-charging system further includes anactuator circuit in communication with the sensor circuit, with theactuator circuit being configured to receive the control signal from thesensor circuit. The actuator circuit is further configured to enable ordisable, based on the control signal, a pre-charge current forpre-charging a selected node of the PA control circuit.

In some embodiments, the sensor circuit can include an op-amp comparatorconfigured to receive the base voltage and the reference voltage asinputs and generate the control signal as an output.

In some embodiments, the actuator circuit can include a switchconfigured to be in an ON state or an OFF state based on the controlsignal to thereby control the enabling or disabling of the pre-chargecurrent. The switch being in the ON state can disable the pre-chargecurrent by shunting the pre-charge current to ground. The switch beingin the OFF state can enable the pre-charge current to be accumulated ina dominant pole capacitance.

In some embodiments, the power amplifier control circuit can include acontrol loop. The control loop can include a dynamic dominant polecircuit configured to generate different dominant poles, with thedynamic dominant pole circuit including the dominant pole capacitanceand a variable dominant pole resistance. The dynamic dominant polecircuit can be configured to generate a first dominant pole by having afirst dominant pole resistance and the dominant pole capacitance, or asecond dominant pole by having a second dominant pole resistance and thedominant pole capacitance. The first dominant pole resistance can behigher than the second dominant pole resistance. The first dominant poleresistance can be selected to yield a first time constant thatfacilitates a stable loop and noise reduction in the control loop, andthe second dominant pole resistance can be selected to yield a secondtime constant that facilitates a fast lock in the control loop.

In some embodiments, the control loop can include a base current sensorconfigured to measure a base current of the PA, and a sensing nodeconfigured to receive the measured base current and a reference currentand perform a current-mode operation to generate an error current. Thecontrol loop can further include a trans-impedance amplifier configuredto amplify the error current to generate an amplified signal provided tothe dynamic dominant pole circuit. The base current sensor can include afinger sensor configured to sense the base current.

According to a number of implementations, the present disclosure relatesto a method for controlling a radio-frequency power amplifier. Themethod includes comparing a base voltage of a power amplifier (PA) and areference voltage selected to be lower than a threshold voltage of thePA. The method further includes generating a control signal based on thecomparison. The method further includes enabling or disabling, based onthe control signal, a pre-charge current for pre-charging a selectednode of the PA control circuit.

In a number of implementations, the present disclosure relates to aradio-frequency (RF) module. The RF module includes a packagingsubstrate configured to receive a plurality of components. The RF modulefurther includes a power amplifier (PA) disposed over the packagingsubstrate. The RF module further includes a PA control circuit disposedover the packaging substrate and interconnected to the PA. The PAcontrol circuit includes a pre-charging system. The pre-charging systemincludes a sensor circuit configured to generate a control signal basedon comparison of a base voltage of the PA and a reference voltageselected to be lower than a threshold voltage of the PA. Thepre-charging system further includes an actuator circuit incommunication with the sensor circuit, with the actuator circuit beingconfigured to receive the control signal from the sensor circuit. Theactuator circuit is further configured to enable or disable, based onthe control signal, a pre-charge current for pre-charging a selectednode of the PA control circuit. RF module further includes a pluralityof connectors configured to provide electrical connections between thepower amplifier, the control circuit, and the packaging substrate.

In some implementations, the present disclosure relates to aradio-frequency (RF) device. The RF device includes a transceiverconfigured to generate an RF signal. The RF device further includes apower amplifier (PA) in communication with the transceiver, with the PAbeing configured to amplify the RF signal. The RF device furtherincludes a PA control circuit in communication with the PA. The PAcontrol circuit includes a pre-charging system. The pre-charging systemincludes a sensor circuit configured to generate a control signal basedon comparison of a base voltage of the PA and a reference voltageselected to be lower than a threshold voltage of the PA. Thepre-charging system further includes an actuator circuit incommunication with the sensor circuit, with the actuator circuit beingconfigured to receive the control signal from the sensor circuit. Theactuator circuit is further configured to enable or disable, based onthe control signal, a pre-charge current for pre-charging a selectednode of the PA control circuit. The RF device further includes anantenna in communication with the PA, with the antenna being configuredto facilitate transmission of the amplified RF signal.

In some embodiments, the RF device can include a wireless device. Such awireless device can be configured to operate as, for example, a GSM/GPRScommunication device.

For purposes of summarizing the disclosure, certain aspects, advantagesand novel features of the inventions have been described herein. It isto be understood that not necessarily all such advantages may beachieved in accordance with any particular embodiment of the invention.Thus, the invention may be embodied or carried out in a manner thatachieves or optimizes one advantage or group of advantages as taughtherein without necessarily achieving other advantages as may be taughtor suggested herein.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 schematically shows a power amplifier (PA) being controlled by acontroller.

FIG. 2 shows a more detailed example of the PA control configuration ofFIG. 1.

FIG. 3 shows additional details of some of the components of thecontroller of FIG. 2.

FIG. 4 shows additional details of a clamping circuit being controlledat least in part by the controller of FIG. 2.

FIG. 5 shows that in some embodiments, the controller of FIG. 2 can beconfigured to operate in a current-mode.

FIG. 6 shows an example of a PA control system having a pre-chargingsystem.

FIG. 7 shows an example of how the pre-charging system of FIG. 6 canprovide improved performance in loop lock acquisition associated withthe PA control system of FIG. 6.

FIG. 8 shows a process that can be implemented to control a PA.

FIG. 9 shows another process that can be implemented to control a PA.

FIG. 10 shows examples of improvements in performance that can beobtained by one or more features of the present disclosure.

FIG. 11 shows that in some embodiments, one or more features of thepresent disclosure can be implemented in a module.

FIG. 12 shows that in some embodiments, the module of FIG. 11 can be apackaged module.

FIG. 13 shows that in some embodiments, the module of FIG. 9 can beimplemented as an integrated front-end module.

FIG. 14 shows that in some embodiments, one or more features of thepresent disclosure can be implemented in a wireless device such as acellular phone.

DETAILED DESCRIPTION OF SOME EMBODIMENTS

The headings provided herein, if any, are for convenience only and donot necessarily affect the scope or meaning of the claimed invention.

In radio-frequency (RF) applications such as wireless power amplifiers(PAs), a current drawn from a power source such as a battery istypically monitored and clamped so as to not exceed a specified valueto, for example, ensure device reliability, prevent excessive heatgeneration, and prolong battery life. It is desirable to have acurrent-limiting technique implemented to ensure that the current drawnfrom the battery does not exceed the specified current limit (e.g.,maximum current limit) while simultaneously not limiting or degradingother performance metrics such as total radiated power (TRP) andswitching transient requirements in normal operation (e.g., voltagestanding wave ratio (VSWR) of 3:1).

In some PA architectures, collector current associated with a PA can besensed directly to facilitate such clamping functionality. In such PAarchitectures, clamping can be facilitated by sensing the collectorcurrent through a small resistance (e.g., a small resistor) andimplementing a clamping circuit that gets triggered by the sensedcurrent value. Another design that utilizes direct sensing of thecollector current includes use of dual bond wires in series with thelast stage of the PA, and a clamping circuit configured to be triggeredbased on the sensed current across the bond wire.

There can be drawbacks associated with direct sensing as a way oflimiting maximum current drawn by a PA. For example, degradation ofefficiency can occur as a resistive sensing element needs to be insertedin series with the PA array. Potential RF corruption of the controllercan also occur. In another example, the direct sensing solution can alsoincrease cost as the sensing element typically needs to be a preciseresistive element. When a precise sensing element is not employed, theaccuracy of the clamping technique is typically not sufficient enough tomeet stringent clamping requirements.

In PA architectures that do not directly sense collector current of aPA, it can become problematic to implement a current-limitingconfiguration to ensure that current drawn from the battery does notexceed a specified amount over extreme conditions. In some of such PAarchitectures, the base bias voltage of the PA can be clamped to notexceed a specified value without any knowledge of the actual current.The lack of knowledge of the actual current value during operation canmake the design of the base bias voltage-based clamp complicated and,more importantly, not effective at limiting current to specified valuein all conditions. An important weakness of the base bias voltage-basedclamping solution can arise from the fact that beta variation of thetransistor (e.g., bipolar junction transistor (BJT)) devices (of the PA)over process and/or temperature is not taken into account. Accordingly,such a clamping design generally proves to be ineffective in limitingcurrent over all conditions.

Disclosed herein are circuits, systems, devices and methods related tocontrolling radio-frequency (RF) power amplifiers (PAs). FIG. 1 shows aPA control configuration 100 where a PA 110 is in communication (e.g.,arrows 122, 124) with a PA control component 120. In someimplementations, such communication can include a feedback loop whereone or more operating parameters associated with the PA 110 is sensed,and based on such sensing, generating a PA control signal to improve theperformance of the PA 110. As described herein, the PA controlconfiguration 100 can include clamping functionalities that can addresssome or all of the foregoing problems.

For the purpose of description herein, a beta parameter β can include aratio between the collector current Icollector and base current Ibase,so that

β=Icollector/Ibase.  (1)

The foregoing relationship in Equation 1 has been shown withexperimental data to generally hold even in extreme conditions thatinclude extreme load mismatch (e.g., VSWR 12:1) and other operatingconditions such as temperature and supply voltages.

In some implementations, a current clamp architecture can be configuredto indirectly sense the current drawn by a PA by utilizing the foregoingbeta parameter relationship. In some implementations, such an indirectsensing technique in conjunction with a technique of generatingbeta-tracking and temperature-compensated reference current can providea way to monitor and/or limit the maximum current drawn by the PA to aspecified value over some or all of operating conditions.

As described herein, a combination that includes some or all of thefollowing can be utilized to clamp a base bias voltage in an effectiveand accurate manner: (i) relationship between the DC collector currentand base current of a PA as given in Equation 1; (ii) a replica of thebase current at the base of the PA; and (iii) a reference current thattracks the variation of beta over process and/or temperature.

FIG. 2 shows an example PA control configuration 100 that can beconfigured to include a clamping architecture having one or morefeatures as described herein. Although described in the example contextof a heterojunction bipolar transistor (HBT) based PA, it will beunderstood that one or more features of the present disclosure can alsobe implemented in PAs that are based on other types of transistors.Also, although described in the example context of a finger-based poweramplifier controller (FB-PAC), it will be understood that one or morefeatures of the present disclosure can be implemented in other types ofPA controllers.

The example configuration 100 is shown to include a PA 110 coupled to aPA controller 120. The PA controller 120 can include a replica basecurrent generator 152 (“Ibase_replica Generator” in FIG. 2) that isconfigured to produce a scaled or unscaled version of the base currentbeing provided to (through path 225) and consumed by the PA 110. Thisreplica base current can be taken to be the PA collector current scaledby its beta.

The PA controller 120 can further include a beta-tracking referencecurrent generator 150 (“beta_tracking Iref Generator” in FIG. 2) that isconfigured to generate a beta-tracking reference current by providing atemperature-compensated bandgap voltage (through path 228) on to a baseresistance (e.g., a base resistor) 132 on the PA 110. The base resistor132 can be configured to track the beta of the BJT process used in thePA design. In some embodiments, the base resistor 132 can be fabricatedfrom the same layer as the base layer of an HBT stack, and can havesimilar characteristic as the base of the BJT devices.

In some embodiments, the base resistor 132 can be configured togenerally track beta inversely over temperature. Thus, the voltageprovided to the base resistor 132 can be temperature-compensated in sucha way that the resulting reference current can be made to track betaover process and temperature.

The replica base current and the beta-tracking reference currentgenerated in the foregoing examples can be utilized to generate areplica base voltage and beta-tracking reference voltage, respectively.For example, replica base current and the beta-tracking referencecurrent can be provided to a matched resistance (e.g., a matchedresistor) to generate the replica base voltage and the beta-trackingreference voltage, respectively.

A differential current steering circuit 154 (“Ibase_clamp” in FIG. 2)can be configured to receive (e.g., through paths 226, 227) and compare(e.g., substantially continuously) the replica base voltage against thebeta-tracking temperature-compensated reference voltage, and steer aproportional current to ground or a clamping node of a PA base driver.In some implementations, a clamping current can be steered into theclamping node of the PA base driver to clamp the PA base bias voltage ifthe replica base voltage exceeds the beta-tracking reference voltage.Additional examples concerning differential current steering circuit 154are described herein in greater detail.

In FIG. 2, the example PA 110 is shown to include an amplifying element130 (e.g., an HBT) that receives the base current Ibase from the PAcontroller 120, through paths 225, 230 and 231. The base of the HBT 130can also receive an RF signal to be amplified (through path 279). Theamplified RF signal is shown to be output from the HBT 130 (e.g.,through the collector) through path 235. The amplified RF signal canthen be provided (through path 237) to a matching circuit 238, aswitching circuit (not shown), and then to an antenna 239 fortransmission.

In the example PA 110, the supply current (Icollector) being provided tothe collector of the HBT 130 from a supply node 236 can be passedthrough, for example, a choke inductance L.

FIG. 2 shows that in some embodiments, paths 225, 230 that provide thebase current Ibase from the PA controller 120 to the PA 110 can be partof a control loop. For example, the base current Ibase provided to thebase of the HBT 130 can also be provided to a sensor 134 such as afinger sensor, through path 232. Examples of such a sensor are describedherein in greater detail.

An output of the sensor 134 (at paths 233, 203) in a current form(Isense) is shown to be combined with a ramp current (Tramp at path202), and the combination (e.g., by direct subtraction of Isense fromIramp), still in a current form, can be amplified by, for example, atrans-impedance amplifier (TIA) 206. A resistance R1 across the TIA 206can be selected to set a gain of the foregoing control loop involvingthe sensor 134. Examples concerning the foregoing current-domainoperation of the sensed current Isense, as well as maintaining areference voltage (Vref) at the sensing node (where Iramp and Isensemeet), are described herein in greater detail.

The output of the TIA 206 is shown to be provided to path 209 and becoupled with the differential current steering circuit 154 (throughpaths 210, resistance R2, paths 211, 124, 215). The resulting voltage atnode 214 combined with the output of the differential current steeringcircuit 154 can be provided to a FET (e.g., a MOSFET) M0 (through path216) for generating a replica base current. The same voltage at path 216is also shown to be provided to the replica base current generator 152through path 234. The output of the TIA 206 is also shown to be coupledto ground at a node after R2, through path 210, capacitance C1, and path213. It is noted that from a controller point of view, the signal ofinterest at the node (where paths 214, 215 and 216 meet) is a voltagewhich gets translated to base voltage that drives the PA. However, froma current clamp point of view, the current steering circuit pushescurrent into the TIA output (209) and to this node, which results indeveloping voltage that overrides the controller voltage and thereforeresults in holding the base voltage of PA at a “clamped” voltage.

The transistor M0 is shown to receive the comparator-output voltage atits gate (through path 216). The source of M0 is shown to be connected(through path 219) to a supply node 267, and the drain of M0 is shown tobe connected (through path 220) to a load resistance RL which is in turngrounded (through path 222). The drain of M0 is also shown to be coupledto the replica base current generator 152 through path 220, 223 and 224.

As described herein, the collector current of the PA can be indirectlysensed, and compared against a beta-tracking temperature-compensatedreference current. Based on such a comparison, the PA base bias voltagecan be clamped so that the total current drawn from a battery does notexceed a specified value in various environmental and device operationconditions.

FIG. 3 shows an example of how the differential current steering circuit154 described in reference to FIG. 2 can be configured. A circuitgenerally indicated as 160 can provide some or all of thefunctionalities associated with the beta-tracking reference currentgenerator 150 of FIG. 2. For example, a reference current that tracksthe beta parameter and is temperature-compensated is shown to begenerated by the circuit 160. More particularly, a reference currentfrom a reference node 267 (through path 258) is shown to be passedthrough a transistor (e.g., a MOSFET) M4 to yield a current in path 259that is provided to the base resistance 132 of the PA 110. Because thebase resistance 132 can track the beta of the HBT 130, the referencecurrent in path 228 can be a beta-tracking reference current.

As shown in FIG. 3, a temperature-compensation feature and the foregoingbeta-tracking feature can be combined to yield a temperature-compensatedbeta-tracking reference voltage. For example, aproportional-to-absolute-temperature (PTAT) current from a bandgapcircuit can be used to generate a temperature-compensated referencevoltage (Vref) at an input of the circuit 160. Then, thistemperature-compensated reference voltage can be dropped across thebeta-tracking base resistance 132 of the PA 110. Thus, Vref, which is afunction of temperature and base resistance (Rb, which is a function ofboth temperature and beta) can be used to generate the reference currentas given by (Vref(temp)/Rb(beta,temp)). Hence, such a reference currentis both temperature compensated and beta-tracking.

The output of the circuit 160 is shown to be provided to the gate of atransistor (e.g., a MOSFET) M3 (through path 260) so as to control theflow of the reference current from the reference node 267 (through path254). M3 and M4 can be configured substantially similarly such that thecurrent exiting M3 in path 255 is a replica of thetemperature-compensated beta-tracking reference current in path 228. Asdescribed herein, such a reference current can be compared against areplica base current by the circuit generally indicated as 162.

In FIG. 3, the foregoing replica base current can be provided to path242 by the circuit generally indicated as 164. The example circuit 164is shown to be implemented as a scaled current mirror configuration,where a reference current from the reference node 267 passes througheach of a first transistor (e.g., a MOSFET) M0 and a second transistor(e.g., a MOSFET) M0/60. Each of the gates of M0 and M0/60 is providedwith a voltage Vint (at node 266), which can be afinger-sensor-compensated reference voltage at path 216 (FIG. 2). In theexample scaled current mirror, the second MOSFET M0/60 is configured toyield a replica base current at path 242 that is scaled from the currentin path 166 by a factor of 60. It will be understood that other scalefactor values can also be utilized.

As shown in FIG. 3, the reference current from M0 is shown to beprovided to the base of an HBT 130 of the PA 110 through paths 166 and168, resistance R3, and path 231.

In FIG. 3, a circuit generally indicated as 162 can provide some or allof the functionalities associated with the differential current steeringcircuit 154 of FIG. 2. The example steering circuit 154 is shown toinclude transistors (e.g., MOSFETs) M1 and M2, with their sourcesreceiving the reference current I1 from the reference node 267 (throughcommon paths 246, 247, and path 248 for M1, and path 250 for M2). Thedrain of M1 is shown to be coupled to ground (through path 249); and thedrain of M2 is shown to provide a clamping current I2 to a clamp node(through paths 251, 215). In some embodiments, M1 and M2 can besubstantially similar so as to allow steering of the reference currentI1 based on comparison of voltages corresponding to the beta-trackingtemperature-compensated reference current in path 255 and the replicabase current in path 242.

The beta-tracking temperature-compensated reference voltage can begenerated by providing the beta-tracking temperature-compensatedreference current to a clamp resistance (Rclamp) through path 256.Similarly, the replica base voltage can be generated by providing thereplica base current to a clamp resistance (Rclamp) through path 243. Insome embodiments, the clamp resistances at their respective paths 256and 243 can be substantially similar so as to allow the foregoingcomparison of the voltages.

The foregoing beta-tracking temperature-compensated reference voltage isshown to be provided to the gate of M2 (through path 245). Similarly,the foregoing replica base voltage is shown to be provided to the gateof M1 (through path 253). Based on such a configuration, thedifferential current steering circuit 162 can compare the foregoingvoltages and steer proportional current to the ground or the clampingnode of a PA base driver.

FIG. 4 shows an example of the foregoing base driver that can beimplemented as a part of a finger-based power amplifier controller(FB-PAC) 120. The current provided to the clamping node from the currentsteering circuit 162 (through path 215) is shown to facilitategeneration of the base bias voltage Vbase by the FB-PAC 120 and providedto the PA through a path indicated as 180. For example, the currentprovided by the current steering circuit 162 (through path 215) can becombined with the output of the op-amp 206 passed through resistance R2to yield the voltage Vint at the clamping node 266. The circuitassociated with the op-amp 206 and related components can be implementedas described herein in reference to FIG. 2.

In FIG. 4, a circuit generally indicated as 269 shows an example of howthe replica base current can be generated from the drain of the scalingtransistor M0/60. As described herein, the source of M0/60 can becoupled to the supply node 267 (through path 241), and the gate of M0/60can be provided with Vint of the clamping node 266 so as to control theflow of the reference current. The current from the drain of M0/60 canbe passed through a transistor (e.g., a MOSFET) M5. The gate of M5 canbe provided with an output of an op-amp 273 (through path 274) thatamplifies the output of M0/60 (through path 271). The op-amp 273 is alsoshown to be supplied with the base bias voltage Vbase. The output ofM0/60 provided to the op-amp 273 can be converted into a voltage by aresistance of 60×RL (through path 270). The resulting current at thedrain of M5 at path 275 is shown to be utilized as the replica basecurrent as described herein.

In some PA control configurations for cellular applications such asGSM/GPRS, a large regulator may be required; and such a regulator can becostly and can impact power added efficiency (PAE). Some PA controlsystems can be configured to eliminate the need for such a largeregulator, to thereby reduce cost and improve PAE. However,sensitivities to RF corruption and DC offset can still presentchallenges. In addition, switching transient (SWT) and power-vs-time(PvT) degradation into voltage standing wave ratio (VSWR) and lack oftotal radiated power (TRP) performance can also suffer.

Some solutions that address the foregoing issues can involve high coststhat prevent widespread adoption for the general GSM/GPRS market. Also,in the example context of GaAs process, some GaAs PA controls can behindered by limitations associated with, for example, GaAs HBT(heterojunction bipolar transistor) process. For example, suchlimitations can include lack of trimming capability, which is typicallyrequired in some PA control circuits.

Some desirable features associated with PAs can include lower costenabled by architecture, superior performance including PAE and TRP,improved robustness and easy-of-use (e.g., an architecture desirablyshould address sensitivity to RF corruption and DC offset, and achieverobust SWT/PvT operation into VSWR), and small module size. Suchdesirable features can apply to GSM/GPRS and other communicationprotocols. In the context of the example GSM/GPRS protocol, a PA's powercontrol architecture can play an important role. Such an architecturecan have profound impact on cost, performance, robustness and modulesize. In some situations, it can be one of the most importantdifferentiating factor in commercial GSM/GPRS PA products. Describedherein are one or more features that can provide a power controlarchitecture that provides advantages in cost, performance, robustness,size, or any combination thereof. In some implementations, all of suchadvantageous features can be effectuated by a power control architecturehaving one or more features described herein.

In some implementations, a power control architecture can include threecomponents: power detection, processing of the detected signal in acontroller, and actuation of a control signal generated by thecontroller. Methods for power detection can include, for example, directdetection and indirection detection. Direct detection typically involvesan RF coupler and RF detectors. Indirect detection can take advantage ofa relationship between RF power and DC characteristics of a given PA.Many modern GSM/GPRS PAs utilize the indirect detection to reduce costand size.

The power actuation referred to above typically includes collectorvoltage control and base current control. Some architectures use acollector voltage of an output stage as a measure of the PA's power.Such an architecture can control the PA power by, for example,regulating the collector voltage through a large low-dropout (LDO). Inmany situations, the large LDO not only increases cost, but it can alsodegrade PAE due to the LDO drop-out.

In some PA control architectures, such as an integrated PA control(IPAC), the need for a large LDO can be eliminated, thereby reducingcost and improving PAE. In addition, a PA can operate in currentsaturation substantially all the time, resulting in improved currentconsumption at different power levels. In such an IPAC architecture, adetected DC current is immediately converted to voltage through aprecision sense resistor. The sense resistor typically needs to be smallin value in order to minimize the voltage drop-out across the resistor.While the small voltage signal minimizes the PAE degradation, it canpresent challenges in the subsequent processing of the signal in thecontroller, due to the sensitivities to RF corruption and DC offsets ofthe error amplifier. To address these issues, some IPAC configurationscan include significant RF bypassing and trimming circuits.

In some embodiments, the power detection referred to above can include amethod where sensing of an output array current is achieved by a sensefinger. Such a sense finger can be configured to closely track theoutput array. The finger current in such a configuration is thenconverted into small voltage through a sense resistor. The voltagesignal is then processed by controller. Additional details concerningthe example sense finger can be found in U.S. Pat. No. 6,734,729,entitled “Closed Loop Power Amplifier Control,” which is expresslyincorporated by reference in its entirety and is to be considered partof the specification of the present application.

In some implementations, the present disclosure relates to a PA controlarchitecture that includes one or more features associated withcurrent-mode control operation. As described herein, such a current-modePA control can be implemented in conjunction with the foregoingfinger-sensing configuration. Although described in such an examplesensing configuration, it will be understood that one or more featuresof the present disclosure can also be implemented in other sensingconfigurations. It will also be understood that one or more features ofthe present disclosure can be implemented in other types of PAs (e.g.,PAs other than HBT, and/or PAs other than those based on GaAs processtechnology).

FIG. 5 shows that in some implementations, a PA control architecture 100can be based on the foregoing finger-based IPAC power detection method.Such an architecture can be similar to or be a substitute of the controlarchitecture examples described herein in reference to FIGS. 2 and 4.

As described herein, a sense finger 532 (e.g., 1×) in a PA circuit 110(e.g., an HBT PA) can be implemented to improve accuracy of tracking anoutput array 530 (e.g., 70×) in DC current. Such a sensing configurationcan include the finger 532 and the output array 530 sharing a common DCballast resistor. The finger 532 can also be placed at a layout locationthat improves thermal tracking. A loading resistance (Rload) to thefinger 532 can be selected with a value that provides compensation ofthe output power's (Pout) dependency on battery voltage (Vbat), therebyimproving power control accuracy across Vbat.

FIG. 5 shows that in some implementations, a finger current (Ifinger) isnot converted into voltage. Instead, analog signal processing of thefinger current in a PA control circuit 120 can remain in the currentdomain. The PA control circuit 120 is shown to include a sensing node500 configured to facilitate such a current mode operation. The sensedfinger current (Ifinger) in path 510 (resulting from the output of thesense finger 532 passed through path 534, resistance Rload, andinductance L2) can be directly subtracted from a reference current(Tramp) in path 508. In the examples described in reference to FIGS. 2and 4, path 510 can be similar to path 203, and path 508 can be similarto path 202.

In some embodiments, the reference current Iramp can be derived from anexternal analog control voltage (e.g., a Vramp signal in a GSM/EDGEsystem). The reference current Iramp can be injected to the sensing node200 through path 508. The sensing node 200 can be regulated tosubstantially maintain at a voltage (Vsense) that tracks Vbat (e.g.,Vsense=Vbat−0.4V).

As further shown in FIG. 5, an error current (Ierror) generated from thesensing node 500 can be provided to (through path 512) and be amplifiedby a trans-impedance amplifier (TIA) 206. In some embodiments, path 512and the TIA 206 can be similar to path 204 and the op-amp 206 in FIGS. 2and 4. A resistance (Rgain, which can be similar to R1 in FIGS. 2 and 4)across the TIA 206 can be selected to set a gain of the example controlloop in FIG. 5. An output voltage of the TIA 206 can be provided to anRF filter (e.g., Rdom and Cdom) through path 514, and then to a basedriver through path 516. The base driver can include a transistor M0which can be similar to M0 in FIGS. 2 and 4. The driver's output can beconnected to the base of the PA as described herein, thereby closing thecontrol loop.

By processing the sensed signal (e.g., finger-sensed signal) in thecurrent domain as described in the foregoing example, the PA controlarchitecture does not need to rely on a high precision sense resistorand/or a high precision error amplifier that requires trimming. Inaddition, the current-mode signal processing can be more robust againstRF corruption, thereby relaxing the requirement for RF bypassing. Forexample, a non-current-mode based IPAC or finger-based IPAC can include3-6 external bypass capacitors to facilitate the power detection andpower control functionalities. In comparison, the example current-basedsensing and control configuration does not require any external bypasscapacitors. In situations where a bypass capacitor is required ordesired, such a capacitor can be integrated in a controller die.

In some implementations, the current-mode processing technique asdescribed herein can provide several advantages. Such a technique caneliminate the need for a high precision sense resistor and/or a highprecision error amplifier, which in turn can eliminate the need fortrimming and allow use of lower cost processes (e.g., a CMOS process) asopposed to more costly processes (e.g., a BiCMOS process). In addition,the current-mode signal processing can significantly improve therobustness against RF corruption, thereby relaxing the requirement forRF bypassing. Accordingly, not requiring components such as externalbypass capacitors can result in a reduction in cost and size of a PAcontrol module.

In some wireless applications (e.g., GSM/GPRS), an output power of apower amplifier can be set by an external voltage reference. Once such areference voltage is applied, it is typically desirable to limitvariations in output power that can occur based on factors such asfrequency, input power, supply voltage, and temperature.

To accomplish such limit in output power variations, a control systemcan be utilized. Such a control system can be based on, for example,analog amplitude control loops. Such loops can involve a number offactors. For example, stability of a loop is typically an importantfactor. To obtain a stable control loop, a dominant pole much below RFfrequencies can be established within the loop. Such a dominant pole canalso be utilized to prevent excessive out of band noise contributionfrom control circuitries.

In other design considerations, it is typically desirable to configure acontrol loop to settle relatively fast before ramping starts, in orderto meet, for example, power-vs.-time (PVT) mask and/or switchingtransient spectrum (SWT) specification. In short, it is typicallydesirable to have the control loop achieve lock quickly, and then rampup to its final value during burst.

In order to lock the loop, an initial constant external referencevoltage can be applied. Typically, such a reference voltage has lowenough power such that it does not violate a given PVT mask. Such avoltage is sometimes referred to as “pedestal” voltage, and thecorresponding power is referred to as “pedestal” power. Similarly, acontrol loop lock time achieved by use of such a pedestal voltage iscommonly referred to as pedestal acquisition time or lock acquisitiontime.

In some implementations, a relatively small dominant pole having a largetime-constant can be utilized to establish a stable loop and to removeor reduce noise associated with the control circuit. On other hand, afast loop lock acquisition is typically achieved with a relatively largedominant pole having a small time-constant. To address such conflictingdesign considerations, an amplitude lock acquisition can be employed toaddress the loop stability and fast loop acquisition design goals.

In systems utilizing such techniques, a dynamic dominant pole andpre-charge methods can be employed to speed up loop acquisition. Inthese systems where loop acquisition is sped up by use of pre-charge anddynamic pole, pre-charge control can play an important role.

In some techniques, a pre-charge system can be controlled with an openloop. In such a system, the loop can be pre-charged to a sufficientlylow value that does not violate design parameters such as PVT mask andforward isolation. Accordingly, such a pre-charge is not fully utilized;and an ideal or desired loop acquisition time may not be obtained.Examples of trade-off between pre-charge duration, PVT mask violation,and loop acquisition time are described herein in greater detail.

FIG. 6 shows an example control system 600 that includes a pre-chargingsystem 620 which can be configured to provide one or more features asdescribed herein. The pre-charging system 620 can be configured with afeedback that charges a desired node (e.g., Vbase) of a PA control loopduring startup to a point (VREF2) just below the PA turn-on threshold sothat forward isolation issue and PVT violation are avoided reduced. As aresult, the loop acquisition can be sped up significantly.

As shown in FIG. 6, the example pre-charging system 620 can include asensor circuit 622 and an actuator circuit 624. The sensor circuit 622can be configured to monitor (e.g., continuously) (through path 616) thebase voltage (Vbase) of the PA 110, and compare it against a selectedreference voltage (VREF2) that is lower than the PA threshold voltage.In some embodiments, such a comparison can be achieved by, for example,an op-amp comparator 614. Based on the comparison of the monitored Vbaseand VREF2, the sensor circuit 622 can generate a control signal at node626.

In some embodiments, the actuator circuit 624 can be in communicationwith the sensor circuit 622 through node 626 so as to receive thecontrol signal generated by the sensor circuit 622. Based on the controlsignal, the actuator circuit 624 can enable or disable a pre-chargecurrent (Iprchrg) that can be utilized to charge up selected node(s) ofthe PA control loop. In some embodiments, such enabling or disabling ofthe pre-charge current (Iprchrg) can be achieved by, for example, aswitch 612 configured to close or open based on the control signal fromthe sensor circuit 622. For example, when the switch 612 is closed, thepre-charge current can be disabled. When the switch 612 is open, thepre-charge current can be enabled.

As shown in FIG. 6, the example pre-charging system 620 can furtherinclude a dynamic pole capability by adjusting the RC value resultingfrom Rdom and Cdom. For example, a resistance (Rdom) associated with adynamic dominant pole can include resistances R4 and R5; and one of R4and R5 can be bypassed (e.g., by a switch 608) when desired to changethe RC value. At low powers, the switch 608 can be closed to short R4,Cdom can be pre-charged to establish good biasing point to speed fastlocking loop. When the switch 608 is opened, Rdom can be increased toincrease the RC value; and such an increase in time constant canfacilitate a stable loop and to remove or reduce noise associated withthe control circuit.

In the example of FIG. 6, the switch 608 can be operated by an output ofa comparator 604 that compares a reference voltage (VREF1) and a rampvoltage (VRAMP). Also in the example of FIG. 6, other portions of the PAcontrol loop (e.g., a current mode sensing node 500, TIA 206, M0(M_DRIVE), and finger sensor of the PA 110) can be configured in similarmanners as described herein in reference to FIGS. 2-5. Althoughdescribed in the context of such an example control loop configuration,it will be understood that one or more features associated with thepre-charging system 620 can be implemented in other types of controlsystems.

The foregoing pre-charging system 620 can be operated to, for example,charge Vbase with a right amount of voltage (e.g., an amount right belowPA turn-on threshold) to thereby better utilize pre-chargingfunctionalities. With the foregoing feedback based pre-chargingconfiguration, the loop lock acquisition is obtained very rapidly forthe PA control loop without impeding other performance metrics.

FIG. 7 shows various plots (as functions of time) obtained by simulationthat demonstrate how the pre-charging system 620 can speed up loopacquisition significantly without degrading PVT performance. In FIG. 7,an example PVT mask specification is indicated by an upper boundary 640and a lower boundary 642. A pre-charging example depicted by curve 644involves an excessive pre-charge (e.g., in an attempt to speed up loopacquisition) which temporarily exceeds the upper boundary 640 therebyviolating the PCT mask specification. To avoid such PVT violation, lesspre-charge can be applied, as shown by curve 646. However, such areduction in pre-charge lengthens the loop acquisition time which maynot meet design specifications.

FIG. 7 further shows a pre-charging curve 648 that can be obtained byuse of the pre-charging system 620 of FIG. 6. One can see that both ofthe PVT mask and loop lock acquisition time specifications are met bysuch a pre-charging curve. As shown and described herein, such anadvantageous feature can result from pre-charging a right amount ofVbase for a right amount of time.

FIG. 8 shows a process 700 that can be implemented to control a poweramplifier (PA). In block 702, a replica base current can be generated,where the replica base current is representative of a collector currentof the PA scaled by a beta parameter. Such a current can be generatedfrom a base current consumed by the PA. In block 704, a beta-trackingreference current can be generated from a temperature-compensatedvoltage (e.g., bandgap voltage) and a base resistor on the PA. In block706, a proportional current can be generated based on the replica basecurrent and the beta-tracking reference current. In block 708, theproportional current can be provided to a clamping node of a PA basedriver under a selected condition (e.g., if the replica base currentexceeds the beta-tracking reference current).

FIG. 9 shows a process 710 that can be implemented to control a poweramplifier (PA). In block 712, a base current of the PA can be measuredto generate a sensed current. In block 714, a current-mode operation canbe performed between a reference current and the sensed current to yieldan error current. For example, the sensed current can be subtracted fromthe reference current to yield the error current. In block 716, acontrol signal can be generated based on the error current. In block718, an operating parameter of the PA can be adjusted based on thecontrol signal. For example, the base current can be adjusted based onthe control signal.

FIG. 10 shows an example of performance that can be achieved by aclamping technique as described herein. The example shows plots ofmaximum current drawn from a battery (Ibatt) through the differentialcurrent steering circuit 154 (“Ibase_clamp” in FIG. 2) at differenttemperature and Vdd, and under what can be considered extreme conditions(e.g., VSWR of 20:1, maximum Vramp (e.g., 2.1V)). The various curves areplotted as functions of phase.

In FIG. 10, an example of maximum allowed specification of 2.4 A isshown. The various Ibatt curves are well below this specified value,with the highest value being approximately 2.25 A. Thus, one can seethat the techniques as described herein can provide an effectiveclamping functionality under a wide range of conditions.

In some implementations, a device and/or a circuit having one or morefeatures described herein can be included in a module. FIG. 11schematically depicts an example module 300, and FIG. 12 shows anexample where such a module can be implemented as a packaged module.

In FIG. 11, the example module 300 is shown to include a PA die 302 thatincludes a plurality of PAs (e.g., HBT PAs) 110. In some embodiments,such PAs can include base resistors for facilitating clamping of basebias voltages as described herein. The PAs 110 are shown to beinterconnected with a PA controller 120 (arrows 122, 124). Suchinterconnections can include one or more features of control loops asdescribed herein. In some embodiments, the PA controller 120 can beimplemented in a die that is separate from the PA die 302. In someembodiments, the PA controller 120 can be implemented in the same die asthe PA die 302.

The module 300 can include connection paths 332, 334, 336 thatfacilitate various operations of the PA controller 120. The connectionpaths 332, 334, 336 can include, for example, connections for providingvarious currents and/or voltages as described herein. The module 300 canalso include other connection paths 330, 338 to facilitate, for example,grounding and other power and/or signals.

In the example module 300, the PA die 302 is shown to include twoexample PAs 110 a, 110 b. However, it will be understood that othernumbers of PA channels can be implemented. In the context of the two PAchannels, the first PA 110 a is shown to be provided with an inputsignal through an input connection 304. Such an input can be passedthrough a matching circuit 306, and an output of the PA 110 a can alsobe passed through a matching circuit 308. The matched output signal canbe output from the module through an output connection 310. Similarly,the second PA 110 b is shown to be provided with an input signal throughan input connection 314. Such an input can be passed through a matchingcircuit 316, and an output of the PA 110 b can also be passed through amatching circuit 318. The matched output signal can be output from themodule through an output connection 320.

In the example packaged module 300 of FIG. 12, a die 302 having a poweramplifier circuit 110 as described herein is shown to be mounted on asubstrate 350. Such a die can be fabricated using a number ofsemiconductor process technologies, including the examples describedherein. The die 302 can include a plurality of electrical contact pads352 configured to allow formation of electrical connections 354 such aswirebonds between the die 302 and contact pads 356 formed on thepackaging substrate 350.

A separate die 360 having a PA controller circuit 120 as describedherein is shown to be mounted on the substrate 350. Such a die can befabricated using a number of semiconductor process technologies,including the examples described herein. The die 360 can include aplurality of electrical contact pads 362 configured to allow formationof electrical connections 364 such as wirebonds between the die 360 andcontact pads 366 formed on the packaging substrate 350.

The packaging substrate 350 can be configured to receive a plurality ofcomponents such as the dies 302, 360 and one or more SMDs (e.g., 380).In some embodiments, the packaging substrate 350 can include a laminatesubstrate.

In the example packaged module 300, a matching circuit 370 can beimplemented on or within the substrate 350. Such a matching circuit 370can include some or all of the match components 306, 308, 316, 318described in reference to FIG. 11.

In some embodiments, the module 300 can also include one or morepackaging structures to, for example, provide protection and facilitateeasier handling of the module 300. Such a packaging structure caninclude an overmold formed over the packaging substrate 350 anddimensioned to substantially encapsulate the various circuits andcomponents thereon.

It will be understood that although the module 300 is described in thecontext of wirebond-based electrical connections, one or more featuresof the present disclosure can also be implemented in other packagingconfigurations, including flip-chip configurations.

FIG. 13 shows that in some embodiments, one or more features of thepresent disclosure can be implemented in a front-end module (FEM) 300configured to facilitate transmit and receive functionalities. Theexample FEM 300 is depicted as providing transmit paths for high and lowbands TX_HB and TX_LB. An RF signal associated with the high band can bereceived at an input pin TX_HB_IN and be provided to a DC-blockcapacitance, and then to a plurality of stages of a first PA 110 a.Similarly, an RF signal associated with the low band can be received atan input pin TX_LB_IN and be provided to a DC-block capacitance, andthen to a plurality of stages of a second PA 110 b. The first and secondPas 110 a, 110 b are shown to be in communication with a control circuit120 having one or more features as described herein. The control circuit120 is shown to be in communication with a plurality of pins forreceiving power as described herein (e.g., Vbatt, Vramp), as well assignals associated with various operations of the control circuit.

In FIG. 13, the control circuit 120 is further shown to provide acontrol signal to a switch circuit 414 configured to provide variousswitching functionalities. For example, TX/RX and band selectionswitching functionalities can be provided by the switch circuit 414.

In FIG. 13, the amplified RF signal from the first PA 110 a is shown tobe routed to the switch circuit 414 through, for example, an outputmatching circuit 308, a DC-block capacitance, and a filter 608.Similarly, the amplified RF signal from the second PA 110 b is shown tobe routed to the switch circuit 414 through, for example, an outputmatching circuit 318, a DC-block capacitance, and a filter 618.

In some embodiments, a front-end module such as the example of FIG. 13can be configured to provide integrated power amplifier controlimplemented in a low profile, compact form factor for quad-band cellularhandsets capable of operating in, for example, GSM850/GSM900 andDCS1800/PCS1900 bands. Such a module can be configured to provide asubstantially complete transmit VCO-to-antenna functionality andantenna-to-receive SAW filter functionality. The example FEM of FIG. 13can also be configured to support Class 12 General Packet Radio Service(GPRS) multi-slot operation and EDGE downlink.

In some implementations, a device and/or a circuit having one or morefeatures described herein can be included in an RF device such as awireless device. Such a device and/or a circuit can be implementeddirectly in the wireless device, in a modular form as described herein,or in some combination thereof. In some embodiments, such a wirelessdevice can include, for example, a cellular phone, a smart-phone, ahand-held wireless device with or without phone functionality, awireless tablet, etc.

FIG. 14 schematically depicts an example wireless device 400 having oneor more advantageous features described herein. In the context of acontrol circuit for a PA, one or more PAs 110 as described herein areshown to be controlled by a PA control circuit 120. Such PAs andcontroller can facilitate, for example, multi-band operation of thewireless device 400. In embodiments where the PAs and their matchingcircuits are packaged into a module, such a module can be represented bya dashed box 300.

The PAs 110 can receive their respective RF signals from a transceiver410 that can be configured and operated in known manners to generate RFsignals to be amplified and transmitted, and to process receivedsignals. The transceiver 410 is shown to interact with a basebandsub-system 408 that is configured to provide conversion between dataand/or voice signals suitable for a user and RF signals suitable for thetransceiver 410. The transceiver 410 is also shown to be connected to apower management component 406 that is configured to manage power forthe operation of the wireless device. Such power management can alsocontrol operations of the baseband sub-system 408 and the module 300.

The baseband sub-system 408 is shown to be connected to a user interface402 to facilitate various input and output of voice and/or data providedto and received from the user. The baseband sub-system 408 can also beconnected to a memory 404 that is configured to store data and/orinstructions to facilitate the operation of the wireless device, and/orto provide storage of information for the user.

In the example wireless device 400, outputs of the PAs 110 are shown tobe matched (via match circuits 420) and routed to an antenna 416 viatheir respective duplexers 412 a-412 d and a band-selection switch 414.The band-selection switch 414 can include, for example, asingle-pole-multiple-throw (e.g., SP4T) switch to allow selection of anoperating band (e.g., Band 2). In some embodiments, each duplexer 412can allow transmit and receive operations to be performed simultaneouslyusing a common antenna (e.g., 416). In FIG. 14, received signals areshown to be routed to “Rx” paths (not shown) that can include, forexample, a low-noise amplifier (LNA).

A number of other wireless device configurations can utilize one or morefeatures described herein. For example, a wireless device does not needto be a multi-band device. In another example, a wireless device caninclude additional antennas such as diversity antenna, and additionalconnectivity features such as Wi-Fi, Bluetooth, and GPS.

Unless the context clearly requires otherwise, throughout thedescription and the claims, the words “comprise,” “comprising,” and thelike are to be construed in an inclusive sense, as opposed to anexclusive or exhaustive sense; that is to say, in the sense of“including, but not limited to.” The word “coupled”, as generally usedherein, refers to two or more elements that may be either directlyconnected, or connected by way of one or more intermediate elements.Additionally, the words “herein,” “above,” “below,” and words of similarimport, when used in this application, shall refer to this applicationas a whole and not to any particular portions of this application. Wherethe context permits, words in the above Description using the singularor plural number may also include the plural or singular numberrespectively. The word “or” in reference to a list of two or more items,that word covers all of the following interpretations of the word: anyof the items in the list, all of the items in the list, and anycombination of the items in the list.

The above detailed description of embodiments of the invention is notintended to be exhaustive or to limit the invention to the precise formdisclosed above. While specific embodiments of, and examples for, theinvention are described above for illustrative purposes, variousequivalent modifications are possible within the scope of the invention,as those skilled in the relevant art will recognize. For example, whileprocesses or blocks are presented in a given order, alternativeembodiments may perform routines having steps, or employ systems havingblocks, in a different order, and some processes or blocks may bedeleted, moved, added, subdivided, combined, and/or modified. Each ofthese processes or blocks may be implemented in a variety of differentways. Also, while processes or blocks are at times shown as beingperformed in series, these processes or blocks may instead be performedin parallel, or may be performed at different times.

The teachings of the invention provided herein can be applied to othersystems, not necessarily the system described above. The elements andacts of the various embodiments described above can be combined toprovide further embodiments.

While some embodiments of the inventions have been described, theseembodiments have been presented by way of example only, and are notintended to limit the scope of the disclosure. Indeed, the novel methodsand systems described herein may be embodied in a variety of otherforms; furthermore, various omissions, substitutions and changes in theform of the methods and systems described herein may be made withoutdeparting from the spirit of the disclosure. The accompanying claims andtheir equivalents are intended to cover such forms or modifications aswould fall within the scope and spirit of the disclosure.

What is claimed is:
 1. A radio-frequency power amplifier controlcircuit, comprising: a sensor configured to measure a base current of apower amplifier and generate a sensed current; a sensing node configuredto receive a reference current and perform a current-mode operation withthe sensed current to yield an error current; and a control loopconfigured to generate a control signal based on the error current toadjust an operating parameter of the power amplifier.
 2. The controlcircuit of claim 1 wherein the power amplifier includes a galliumarsenide (GaAs) heterojunction bipolar transistor (HBT) power amplifier.3. The control circuit of claim 2 wherein the GaAs HBT power amplifieris configured to operate in GSM/GPRS communication protocols.
 4. Thecontrol circuit of claim 1 wherein the sensor includes a finger sensorconfigured to sense the base current of the power amplifier to yield thesensed current.
 5. The control circuit of claim 1 wherein thecurrent-mode operation includes subtracting the sensed current from thereference current to yield the error current.
 6. The control circuit ofclaim 1 wherein the power amplifier control circuit is substantiallyfree of a sense resistor between the sensor and the sensing node.
 7. Thecontrol circuit of claim 1 wherein the reference current is obtainedfrom an external analog control voltage.
 8. The control circuit of claim1 wherein the sensing node is configured to be regulated tosubstantially maintain a voltage that tracks a battery voltage.
 9. Thecontrol circuit of claim 1 wherein the control loop includes atrans-impedance amplifier configured to amplify the error current. 10.The control circuit of claim 1 wherein the control loop includes aclosed control loop.
 11. The control circuit of claim 1 wherein thepower amplifier control circuit is substantially free of an externalbypass capacitor to thereby reduce cost and size associated with thepower amplifier circuit.
 12. The control circuit of claim 1 furthercomprising a pre-charging system configured to pre-charge a selectednode of the control circuit.
 13. The control circuit of claim 12 whereinthe pre-charging system including a sensor circuit configured togenerate a control signal based on comparison of a base voltage of thePA and a reference voltage selected to be lower than a threshold voltageof the PA, the pre-charging system further including an actuator circuitin communication with the sensor circuit, the actuator circuitconfigured to receive the control signal from the sensor circuit, theactuator circuit further configured to enable or disable, based on thecontrol signal, a pre-charge current for pre-charging the selected nodeof the control circuit.
 14. A method for controlling a radio-frequencypower amplifier, the method comprising: measuring a base current of apower amplifier to generate a sensed current; performing a current-modeoperation between a reference current and the sensed current to yield anerror current; generating a control signal based on the error current;and adjusting an operating parameter of the power amplifier based on thecontrol signal.
 15. The method of claim 14 wherein performing thecurrent-mode operation includes subtracting the sensed current from thereference current to yield the error current.
 16. The method of claim 14wherein generating the control signal includes amplifying the errorcurrent with a trans-impedance amplifier.
 17. The method of claim 14wherein at least some of the generating and adjusting are facilitated bya closed control loop.
 18. A radio-frequency (RF) module comprising: apackaging substrate configured to receive a plurality of components; apower amplifier (PA) disposed over the packaging substrate; a controlcircuit disposed over the packaging substrate and interconnected to thePA, the control circuit including a sensor configured to measure a basecurrent of the PA and generate a sensed current, a sensing nodeconfigured to receive a reference current and perform a current-modeoperation with the sensed current to yield an error current, and acontrol loop configured to generate a control signal based on the errorcurrent to adjust an operating parameter of the PA; and a plurality ofconnectors configured to provide electrical connections between the PA,the control circuit, and the packaging substrate.
 19. The RF module ofclaim 18 wherein the PA is disposed on a first die and the controlcircuit is disposed on a second die, each of the first die and thesecond die being mounted on the packaging substrate.
 20. The RF moduleof claim 18 wherein the control circuit further includes a pre-chargingsystem configured to pre-charge a selected node of the control circuit,the pre-charging system including a sensor circuit configured togenerate a control signal based on comparison of a base voltage of thePA and a reference voltage selected to be lower than a threshold voltageof the PA, the pre-charging system further including an actuator circuitin communication with the sensor circuit, the actuator circuitconfigured to receive the control signal from the sensor circuit andenable or disable, based on the control signal, a pre-charge current forpre-charging the selected node of the control circuit.